Transmission process having spectrum spread phase differential modulation adn demodulation using orthogonal pseudorandom sequences

ABSTRACT

Transmission process having spectrum spread phase differential modulation and demodulation using orthogonal pseudorandom sequences. Use is made of two orthogonal pseudorandom sequences (C E1 , C E2 ) for spreading the spectrum of the symbols (d k ) to be transmitted and detection takes place of the phase difference between the received signals filtered by filters matched to the two sequences, one being delayed.

TECHNICAL FIELD

The present invention relates to a transmission process having spectrumspread phase differential modulation and demodulation using orthogonalpseudorandom sequences.

The direct sequence spectrum spread modulation method has already beenused for many years, more particularly in radiocommunications withsatellites and in the military field. It is also used in radiolocation,such as in the so-called global positioning system or GPS.

This method has numerous advantages:

a) Discretion: This discretion is linked with the spread of theinformation transmitted over a broad frequency band. This leads to a lowspectral density of the transmitted power. This feature makes thismethod particularly attractive for military applications, which requiremaximum discretion transmission systems. It is also of interest forcivil applications, because it makes it possible to allocate the samefrequency band to services using conventional spectrum spread modulationmethods.

b) The resistance to scramblers and interference: A spectrum spreadreceiver effects the correlation operation between the message receivedand the spread pseudorandom sequence before any demodulation operation.This correlation operation makes it possible to extract the message tobe received and rejects any scrambler or interference. The longer thepseudorandom sequence used for spreading the spectrum the greater theresistance to scramblers. This feature is vital in militaryapplications. It is also useful in civil applications, because it isnecessary to avoid, apart from interference due to users operating in anarrow band, the broad band interference produced by other spectrumspread transmissions.

c) The excellent behaviour in fading channels: Provided that use is madeof an asynchronous receiver structure, this method makes it possible totransmit digital data in reliable manner between mobiles and in thepresence of obstacles generating multiple propagation paths. This is theconsequence of the diversity effect inherent in the system making itpossible to demodulate independently of one another all the informationscarried by all the propagation paths. This feature makes it possible toenvisage the use of this method for all transmissions between vehiclesin an urban environment, or for radio connections within buildings,where fading effects can be produced by the presence of personnelconstituting mobile obstacles for radio waves.

d) The possibility of using a multiple access protocol with distributionby codes: This method consists of allocating orthogonal spreadpseudorandom sequences (i.e. having low intercorrelation coefficients)to the different users. However, it remains difficult to implement,because it imposes an effective control of the transmission power.

The direct sequence spectrum spread modulation method has been describedto a significant extent in the specialized literature. Reference can bemade to two works, namely "Spread spectrum communications" by Marvin K.Simon et al, published by Computer Science Press, USA and the secondentitled "Spread spectrum systems" by Robert C. Dixon and published byJohn Wiley and Sons, USA.

PRIOR ART

In a conventional data transmitter using a conventional modulationmethod, modulation takes place of a radio frequency carrier by phase,frequency or amplitude modulation, or by a mixed modulation. Forsimplification purposes reference will only be made to phasemodulations, which are nowadays the most frequency used. The digitaldata to be transmitted are binary elements or bits, which have a periodT_(b), i.e. a new bit must be transmitted for every T_(b). With thesebits it is possible to constitute symbols having a period T_(s). Theseare symbols which will modulate the radio frequency carrier. The symbolflow rate is expressed in bauds (or symbols/second).

This procedure can be illustrated by two examples:

a) Binary phase shift keying (BPSK), which consists of allocating aphase state, e.g. 0, to the 0 bits and another phase state, e.g. π tothe 1 bits. In this case, the symbol is the bit and T_(s) =T_(b). Thephase state of the radio frequency carrier is imposed for every bit.

b) Quaternary phase shift keying or QPSK consists of using symbolsformed by two successive bits. Thus, these symbols can assumefour-states (00, 01, 10, 11). A phase state of the carrier is allocatedto each of these states and in this case T_(s) =2T_(b). Therefore thephase state of the radio frequency carrier is imposed every other bit.

It is possible to improve the transmitted radio signal spectrum and inparticular reduce the power in parasitic side lobes by using moresophisticated phase modulations, for which the modulating signal isshaped (filtered) prior to modulation and reference is then made to MSK,GMSK, SRC4 modulations.

On the reception side a distinction can be made between two largedemodulation families: coherent demodulation and non-coherentdemodulation. The coherent demodulation methods consists of implementingin the receiver a subassembly, whose function is to estimate the averagephase of the carrier, so as to reconstitute a phase reference, which isthen mixed with the signal received in order to demodulate the data.

This method has the optimum performance characteristics in radiochannels, where a Gaussian noise is added to the useful signal. This ise.g. the case with transmissions with satellites. However, in thepresence of multiple paths, this method gives very poor results.

The non-coherent demodulation method is based on the observationaccording to which it is sufficient for the phase reference of thecurrent symbol is the phase of the preceding symbol. In this case, thereceiver does not estimate the phase of the symbols, but instead thephase difference between two successive symbols. Thus, there isdifferential phase shift keying (DPSK) or differential quadrature phaseshift keying (DQPSK).

FIG. 1 shows the block diagram of a DPSK transmitter, which has an inputEe, which receives the data b_(k) to be transmitted, which have a periodT_(b) and it comprises a differential coder 10 formed by a logic circuit12 and a delay circuit 14, a local oscillator 16 supplying a carrier Pand a modulator 18 connected to an output Se supplying the DPSK signal.

The logic circuit 12 receives the binary data b_(k) and supplies otherbinary data called symbols and designed d_(k). The logic circuit 12 alsoreceives the symbols delayed by a period T_(b), i.e. d_(k-1). The logicoperation performed in the circuit 12 is the exclusive-OR operation onthe data b_(k) and on the compliment of d_(k) delayed by one serialposition (i.e. d_(k-1) ), which gives: d_(k) =b_(k) .sup.⊕ d_(k-1) .

The modulator 18 phase modulates the carrier P as a function of eachsymbol d_(k).

FIG. 2 shows the block diagram of a corresponding receiver of thedifferential demodulator type. This receiver has an input Er andcomprises a band pass filter 20, a delay circuit 22 for a duration Tb, amultiplier 24, an integrator 26 operating on a period Tb and a logicdecision circuit 28. The receiver has an output Sr, which restores thedata b_(k).

The pass band of the input filter 20 is between the NYQUIST band equalto 1/T_(b) and the width of the main lobe of the DPSK signal, which is2/T_(b).

By designating x(t) the signal received applied to the input Er, themultiplier 24 received a filtered signal x_(F) (t) emanating directlyfrom the filter 20 and a delayed filtered signal x_(FR) emanating fromthe delay circuit 22. It supplies a product signal m(t), which isintegrated on a period T_(b) in the integrator 26, which supplies asignal, whose polarity makes it possible to determine the value of thetransmitted bit, which is decided by the logic 28.

The direct sequence spectrum spreading method consists of multiplyingthe sequence of symbols d_(k) obtained after the differential coding bya pseudorandom sequence having a much higher flow rate than that of thedata to be transmitted. This pseudorandom sequence has a binary rate Ntimes higher than that of the binary data to be transmitted. Theduration T_(c) of one bit of this pseudorandom sequence, said bit alsobeing called a chip, is therefore equal to T_(b) /N. The chip rate ofthe pseudorandom sequence can be several megachips, or even severaldozen megachips per second.

FIG. 3 shows the block diagram of a direct sequence spectrum spreadtransmitter. The elements already shown in FIG. 1 carry the samereferences. In addition to those of FIG. 1, the emitter comprises apseudorandom (C) sequence generator 30 and a multiplier 32. The signalsupplied by the multiplier has a pseudorandom sequence-spread spectrum.The modulator 18 then no longer operates on the symbol of origin d_(k),but on the corresponding spread symbol D_(k).

The corresponding receiver has the same structure as that of FIG. 2,except that the filter 20 is now a matched filter, whose pulse responsereduced to the baseband is the time-reversed combined complex of thepseudorandom sequence used in the transmitter.

The pseudorandom sequence used on transmission must consequently have anautocorrelation function with a marked peak of value N for a zero delayand the smallest possible side lobes. This can be obtained by usingmaximum length sequences also known as m-sequences, or so-called GOLD orKASAMI sequences.

The sequences usable in a spectrum spread system have been the subjectmatter of intense studies, which are summarized in the article entitled"Cross-correlation properties of pseudorandom and related sequences" byDilip V. SARWATE and Michael B. PURSLEY, published in "Proceedings ofthe IEEE", vol. 68, No. 5, May 1980, pp 593 to 619.

The input filter used in a spectrum spread receiver has an equivalentbaseband pulse response H(t) and this response must be the time-reversedcombined complex of the pseudorandom sequence used on transmission:

    H(t)=c*(T.sub.s -t)

where C(t) is the pseudorandom sequence used on transmission.

Therefore the signal supplied by such a filter is:

    x.sub.F (t)=x(t)*H(t)

where the symbol * designates the convolution operation, so as to give:##EQU1##

Thus, the matched filter performs the correlation between the signalapplied to its input and the pseudorandom spread sequence.

In a Gaussian additive noise channel, the signal x_(F) (t) willtherefore be in the form of a pulse signal, whose repetition frequencyis equal to 1/T_(b). The envelope of this signal is the autocorrelationfunction of the signal C(t). The information is carried by the phasedifference between two successive correlation peaks. Thus, themultiplier output will be formed by a succession of positive or negativepeaks, as a function of the transmitted bit value.

In the case of a radiotransmission in the presence of multiple paths,the output of the matched filter will be formed by a succession ofcorrelation peaks, each peak corresponding to a propagation path.

The different signals of the reception chain are shown in FIG. 4. Theline (a) represents the filtered signal x_(F) (t), the line (b) thecorrelation signal x_(F) (t)*x_(F) (T_(s) -t) and the line (c) thesignal at the integrator output.

This known procedure is described in detail in the article entitled"Direct-Sequence Spread Spectrum with DPSK Modulation and Diversity forIndoor Wireless Communications" published by Mohsen KAVEHRAD and BhaskarRAMAMURTHI in "IEEE Transactions on Communications", vol. COM 35, No. 2,February 1987.

Thus, a description has been given of DPSK spectrum spread modulation.It is clear that this procedure can be applied in the same way to alldifferential modulations. For example, in the case of DQPSK modulation,each correlation peak at the output of the matched filter could assumefour phase states, whereas there are only two with a DPSK modulation.

Despite their great interest these procedures suffer from disadvantages.Thus, the successive bits constituting the message to be transmitted areall multiplied by the same pseudorandom spread sequence, which can giverise to a coherent intercorrelation noise able to deteriorate thequality of the signal m(t) at the demodulator output.

The applicant has profoundly studied this question and the researchcarried out can be summarized as follows. Consideration must again begiven to the transmitter shown in FIG. 3 and the receiver of FIG. 2,knowing that the latter operates with a matched filter 20 correspondingto a given pseudorandom sequence. In general terms, the sequence used onreception is designated C_(R) and that used on transmission C_(E). Whenthe transmitter and receiver are matched to one another, one obviouslyobtains C_(E) .tbd.C_(R).

In baseband, the signal at the input of the matched filter 20 can bewritten:

    x(t)=dC.sub.E (t)

d representing the sequence of bits modulating the carrier, said bitsbeing obtained from binary data to be transmitted b following passage inthe differential coder 10.

The filtered signal found at the output of the matched filter can thenbe written: ##EQU2##

C_(R) * representing the combined complex of C_(R), which can then bewritten: ##EQU3## or once again

    x(t)=d.sub.-1 C.sub.C.sbsb.E.sub.,C.sbsb.R (-t)+d.sub.0 C.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t)

where the notation C_(C).sbsb.E.sub.,C.sbsb.R represents the aperiodiccorrelation function between C_(E) and C_(R).

In the case where C_(E) .tbd.C_(R) the same formula is obtained byreplacing the C_(C).sbsb.E.sub.,C.sbsb.R by C_(C).sbsb.R, which is theaperiodic autocorrelation function of C_(R).

The delayed filtered signal found at the output of the delay circuit 22is simply shifted by one symbol and can consequently be written:

    X.sub.FR (t)=d.sub.-2 C.sub.C.sbsb.E.sub.,C.sbsb.R (-t)+d.sub.-1 C.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t).

Finally, at the output of the demodulator 24, the signal can be written:

    m(t)=x.sub.F (t).x.sub.FR (t)

or

    m(t)=d.sub.-2 d.sub.-1 C.sup.2.sub.C.sbsb.E.sub.,C.sbsb.R (-t)+{d.sup.2.sub.-1 +d.sub.-2 d.sub.0 }C.sub.C.sbsb.E.sub.,C.sbsb.R (-t).C.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t)+d.sub.-1 d.sub.0 C.sup.2.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t)

As the symbols d are obtained from data b by the relation d_(k) =b_(k)⊕d_(k-1) and as d or b can only assume the two values +1 or -1, wenecessarily obtain:

    d.sub.-2 d.sub.-1 =b.sub.-1, d.sub.-1.d.sub.0 =b.sub.0

and

    d.sup.2.sub.-1 +d.sub.-2.d.sub.0 =1+b.sub.-1.b.sub.0

Thus, one once again obtains a new expression of the demodulation signalm(t):

    m(t)=b.sub.-1 C.sup.2.sub.C.sbsb.E.sub.,C.sbsb.R (-t)+{1+b.sub.0 b.sub.-1 }C.sub.C.sbsb.E.sub.,C.sbsb.R (-t)C.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t)+b.sub.0 C.sup.2.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -(t)

Three cases can then occur as a function of the respective values of b₀and b₋₁.

Case No. 1: b₀ =b₋₁ =1

We then obtain:

    m(t)={C.sub.C.sbsb.E.sub.,C.sbsb.R (-t)+C.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t)}.sup.2 =⊖.sup.2.sub.C.sbsb.E,.spsb.C.sbsb.R (T.sub.b -t)

in which ⊖² _(C).sbsb.E.sub.,C.sbsb.R is called the even correlationfunction of C_(E) and C_(R).

Case No. 2: b₋₁ =b₀ =-1

    m(t)=-{C.sub.C.sbsb.E.sub.,C.sbsb.R (-t)-C.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t)}.sup.2 =-⊖.sup.2.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b-t)

where ⊖² _(C).sbsb.E.sub.,C.sbsb.R is called the odd autocorrelationfunction of C_(E) and C_(R).

Case No. 3: b₋₁ =-b₀

    m(t)=b.sub.0 {C.sup.2.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t)-C.sup.2.sub.C.sbsb.E.sub.,C.sbsb.R (-t)}=b.sub.0 ⊖.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t).⊖.sub.C.sbsb.E.sub.,C.sbsb.R (T.sub.b -t)

With these considerations developed, the deficiency of the conventionaldirect sequence spectrum spread method becomes very clear. Thus, atleast in the first two cases, the demodulation signal m(t) is eitheralways positive, or always negative. However, it is this signal whichwill be integrated into the circuit 26 prior to decision taking. Apartfrom the correlation peak obtained for t=0, this signal should ideallyhave a zero mean value. This is manifestly not the case and theintegrator output will consequently have at the time of decision making,a significant negative or positive shift value as a result of theintegration of a signal, whose mean value is not zero. These shifts willdeteriorate the quality of the decision taking and will increase the biterror rate and this will become worse as the integration time increases.

The object of the invention is to obviate this disadvantage and aims atimproving the quality of the output signal of the demodulator (signalm(t)) and therefore the transmission quality.

DESCRIPTION OF THE INVENTION

The invention therefore proposes a novel transmission process byspectrum spread phase differential modulation and demodulation in whichuse is made of two pseudorandom sequences on transmission, which areorthogonal to one another, said two sequences being used for spreadingthe spectrum of the signals, which will then modulate a carrier. Onreception, two filtering operations matched to these two orthogonalsequences will take place. In order to obtain the demodulation signal,multiplication will take place of the delayed filtered signal associatedwith one of the sequences by the filtered, but not delayed signalassociated with the other sequence. As these two signals result fromfiltering operations by filters, whose pulse responses are orthogonal inthe NYQUIST sense, the interference affecting these signals will bedecorrelated, which will decrease their influence on the demodulationsignal. In the prior art, the two signals, respectively delayed andfiltered and filtered emanated from the same filter, so that theinterference affecting them was correlated.

More specifically, the present invention relates to a data transmissionprocess by spectrum spread phase differential modulation anddemodulation, in which the data are organized in at least one binarydata sequence having a certain period, characterized in that:

A) on transmission:

a differential coding of the binary signals to be transmitted takesplace in order to obtain symbols, use being made of two orthogonalpseudorandom sequences for spreading the spectrum of each symbol and acarrier is modulated by the spread spectrum symbols,

B) on reception:

two parallel filtering operations take place matched to the twoorthogonal sequences and there is an estimation of the phase differencebetween the two signals obtained after filtering, one of which has beendelayed, which restores the data.

The invention has several embodiments according to which transmissiontakes place of one, two or more than two data flows.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1, already described, shows the block diagram of a prior art,differential coding transmitter.

FIG. 2, already described, shows the block diagram of a correspondingreceiver.

FIG. 3, already described, shows the block diagram of a prior art,direct sequence spectrum spread transmitter.

FIG. 4, already described, shows the configuration of the differentsignals appearing in a direct sequence spectrum spread receiver.

FIG. 5 is a block diagram of a transmitter according to the invention ina first variant with a single flow of data to be transmitted.

FIG. 6 is a block diagram of a corresponding receiver.

FIG. 7 symbolically shows the transmission principle of a single dataflow.

FIG. 8 symbolically shows the principle of simultaneously transmittingtwo data flows.

FIG. 9 shows the structure of a transmitter corresponding to a doubledata transmission.

FIG. 10 shows the structure of the corresponding receiver.

FIG. 11 symbolically illustrates a transmission of three data flows withthree pairwise orthogonal pseudorandom sequences.

FIG. 12 illustrates a transmission variant of two symbols with the halfflow rates.

FIG. 13 illustrates an embodiment of a receiver having a surfaceacoustic wave component.

DETAILED DESCRIPTION OF EMBODIMENTS

FIG. 5 shows a diagram of an embodiment of a transmitter according tothe invention. This transmitter comprises the means already shown inFIG. 3 and which carry the same references, namely an input Ee receivingthe binary data b_(k) to be transmitted, said data having a periodT_(b), a differential coder 10 with its logic circuit 12 and its delaycircuit 14. This coder supplies the symbols d_(k). The transmitter shownalso comprises a first generator 30₁ supplying a first pseudorandomsequence C_(E1) applied to a first multiplier 32₁ and a second generator30₂ supplying a second pseudorandom sequence C_(E2) applied to a secondmultiplier 32₂. The multipliers 32₁ and 32₂ supply symbols (D_(k))₁ and(D_(k))₂, whose spectrum is spread by the corresponding sequences. Thetransmitter also comprises a first modulator 18₁ receiving a carrier Pproduced by a local oscillator 16 and the symbols (D_(k))₁ spread by thefirst pseudorandom sequence C_(E1) and a second modulator 18₂ receivingsaid same carrier P and the symbols (D_(k))2 spread by the secondpseudorandom sequence C_(E2). Finally, an adder 31 is connected to thetwo modulators 18₁ and 18₂ and supplies the output Se of thetransmitter. This output transmits the carrier modulated by the doublyspread signal. The modulation can be of a random nature (phase,frequency, mixed, etc.).

Naturally, it would be possible to modify this circuit by firstly addingthe two spread signals (D_(k))₁ and (D_(k))₂ and by using a singlemodulator. It would also be possible to add the two pseudorandomsequences C_(E1) and C_(E2) as soon as they are formed and to spread thesignal with the aid of the pseudorandom sum obtained.

The receiver shown in FIG. 6 comprises the means already shown in FIG.2, namely an input Er receiving a signal x(t), a delay circuit 22 with aduration equal to the period T_(b) of the transmitted bits, a multiplier24 supplying a demodulation signal m(t), an integrating circuit 26acting on the signal m(t) supplied by the multiplier 24, the integrationtaking place between 0 and T_(b), a logic decision circuit 28 and ageneral output Sr supplying the original digital data b_(k). Thisreceiver differs from the prior art in that it comprises a first filter20₁ matched to the first pseudorandom sequence C_(E1) used in thetransmitter and a second filter 20₂ matched to the second pseudorandomsequence C_(E2) used in the transmitter. The term filter matched to apseudorandom sequence is understood to mean a filter, whose pulseresponse brought down to the baseband is the time-reversed combinedcomplex of the pseudorandom sequence in question.

The first filter 20₁ supplies a filtered signal x_(F1), which is thendelayed by T_(b) in a circuit 22 in order to give a delayed, filteredsignal x_(FR1). The second filter 20₂ supplies a filtered signal x_(F2).This signal is not delayed and is used for producing the demodulationsignal m(t) by multiplication with the filtered, delayed signal x_(FR1).

This transmitter-receiver assembly functions as follows. As in the caseof conventional modulation with direct sequence spectrum spreading, thebinary quantities used for modulating the carrier are the symbols d_(k)obtained from the original data b_(k) following passage in thedifferential coder 10. In the conventional process, the symbols d_(k)are multiplied by a single pseudorandom sequence C_(E). In the processaccording to the invention, the symbols d_(k) are multiplied by the twopseudorandom sequences C_(E1) and C_(E2), which are orthogonal to oneanother, in order to form a signal which will modulate the carrier.

The sequences C_(E1) and C_(E2) can have the same generalcharacteristics as a sequence used in the prior art. They are bothconstituted by N elements or chips having a period T_(c) =T_(b) /N andtheir autocorrelation functions have a single marked peak for a zerodelay. They have the lowest possible value elsewhere. These sequencesare also chosen so as to have a low intercorrelation function, no matterwhat the time.

In the case of the receiver, the input signal x(t) can be written:

    x(t)=d(C.sub.E1 (t)+C.sub.E2 (t))

The filtered, delayed signal x_(FR1) at the output of the delay circuit22 can be written:

    x.sub.FR1 =d.sub.-2 {C.sub.CE1,CE1 (-t)+C.sub.CE2,CE1 (-t)}+d.sub.-1 {C.sub.CE1,CE1 (T.sub.b -t)+C.sub.CE2,CE1 (T.sub.b -t}

In the same way, the filtered signal X_(F2) at the output of the filter20₂ can be written:

    x.sub.F2 =d.sub.-1 {C.sub.CE1,CE2 (-t)+C.sub.CE2,CE2 (-t)}+d.sub.0 {C.sub.CE1,CE2 (T.sub.b -t)+C.sub.CE2,CE2 (T.sub.b -t)}

At the output of the multiplier 24, the demodulation signal is of form:

    m(t)=x.sub.FR1.x.sub.F2

Therefore four cases can be distinguished:

Case No. 1: b₀ =b₋₁ =1

We then have d₋₂ =d₋₁ =d₀. It is then possible to rewrite x_(FR1) andX_(F2) in the following form:

    x.sub.FR1 =d.sub.-1 {⊖.sub.CE1 (T.sub.b -t)+⊖.sub.CE2,CE1 (T.sub.b -t)}

and

    x.sub.F2 =d.sub.-1 {⊖.sub.CE1,CE2 (T.sub.b -t)+⊖.sub.CE2 (T.sub.b -t)}

where, after simplification and replacing ⊖_(CEi),CEj (T_(B) -t) with⊖_(ij)

    m(t)=⊖.sub.11.⊖.sub.12 +⊖.sub.11.⊖.sub.22 +⊖.sub.21.⊖.sub.12 +⊖.sub.21.⊖.sub.22

For t=T_(b), we clearly obtained m(0)=⊖₁₁ (0).⊖₂₂ (0)+ . . . i.e. N²,which corresponds to the autocorrelation peak.

A detailed analysis of the signal m(t) makes it possible to see that inthe case where the two sequences used for modulating the data areorthogonal, the signal m(t) has a zero mean value outside thecorrelation peak. Thus, in this case, the output of the integrator 26does not have a shift as was the case with the conventional structure.

Case No. 2: b₀ =b₋₁ =-1

In this case, we obtain d₀ =d₋₁ =-d₋₂. Thus, it is possible to rewritex_(FR1) and x_(F2) in the following form, with the same simplifyingnotations as hereinbefore:

    X.sub.FR1 -d.sub.-1 {⊖.sub.11 +⊖.sub.21 }

    x.sub.F2 =-d.sub.-1 {⊖.sub.12 +⊖.sub.22 }

so that m(t)=⊖₁₁.⊖₁₂ +⊖₁₁.⊖₂₂ +⊖₂₁.⊖₁₂ +⊖₂₁.⊖₂₂

All the remarks made in connection with case No. 1 still apply.

Case No. 3: b₀ =1 b₋₁ =-1

In this case, we obtain d₀ =d₋₁ =-d₋₂ and therefore

    x.sub.FR1 =d.sub.-1 {⊖.sub.11 +⊖.sub.21 }

    x.sub.F2 =d.sub.-1 {⊖.sub.12 +⊖.sub.22 }

and m(t)=⊖₁₁.⊖₁₂ +⊖₂₁.⊖₁₂ +⊖₁₁.⊖₂₂ +⊖₂₁.⊖₂₂

The same remarks still apply.

Case No. 4: b₀ =-1 b₋₁ =1

In this last case, we have d₀ =-d₋₁ =-d₋₂ and

    x.sub.FR1 =d.sub.-1 {⊖.sub.11 +⊖.sub.21 }

    x.sub.F2 =-d.sub.-1 {⊖.sub.12 +⊖.sub.22 }

so that m(t)=-⊖₁₁ ⊖₁₂ -⊖₁₁ ⊖₂₂ -⊖₂₁ ⊖₁₂ -⊖₂₁ ⊖₂₂

The same remarks still apply.

The two signals received by the demodulator 24 are different both asrespect to their serial position and by the pseudorandom sequenceinvolved. The signal x_(FR1) has been delayed by a period T_(b) andtherefore corresponds, on transmission, to a symbol of serial positionk-1, whereas the signal x_(F2) corresponds to the following symbol ofserial position k. In addition, the signal x_(FR1) comes from a symbolwhich has been spread and then despread by the first sequence C_(E1),whereas the signal x_(F2) has been processed by the second sequenceC_(E2). As a result of the delay of one serial position between thesymbols in question, it can be seen that the sequences C_(E1) and C_(E2)have not been transmitted at the same time. Each sequence isperiodically transmitted with the same period T_(b) as the symbols to beprocessed, but the sequence C_(E1) linked wit the signal x_(FR1) hasbeen transmitted one serial position prior to the sequence C_(E2) linkedwith the signal x_(F2). The two sequences C_(E1) and C_(E2), althoughglobally produced simultaneously by the generators 30₁ and 30₂,intervene in a manner shifted by one serial position as a result of theprocessing performed on reception of the sequences. Thus, use is thenmade of a sequence C_(E1) and then a sequence C_(E2).

This process is diagrammatically illustrated in FIG. 7. The firstsequence C_(E1) is repeated periodically for each symbol d_(k-1), d_(k),d_(k+1), etc., as is the second sequence C_(E2). These sequences aresymbolized by the rectangles of respectively the upper and lower bands.On reception, the demodulation signal m(t) is formed from a signalx_(FR1), whose origin is a symbol of serial position k-1 processed byC_(E1) and a signal x_(F1) originating from a symbol of serial positionk processed by C_(E2). The downward oblique arrow from a sequence C_(E1)of serial position k-1 to a sequence C_(E2) of serial position ksymbolizes the chronology of the sequences used. This chronology isrepeated on each formation of a new demodulation signal. All thedownward oblique arrows symbolize the transmission of all the symbolsd_(k), when k increases unitwise.

In the embodiment illustrated in FIGS. 5 and 6, this distinction betweenthe serial position of the sequences involved has no importance becausea sequence is identically repeated (the sequence C_(E1) of serialposition k-1 being identical to the sequence C_(E1) of serial positionk). However, the distinctions described hereinbefore give a betterunderstanding of the improvement provided by the invention and which isas follows.

It is possible to conceive a receiver using a first matched filter onC_(E1) and which would not be followed by a delay circuit, the latterbeing associated with the second filter matched to C_(E2). Demodulationwould then take place of the phase jumps caused by C_(E2) followed byC_(E1).

In other words, it is possible to exploit the presence of twopseudorandom sequences on transmission by simultaneously transmittingtwo data flows and by detecting the phase jumps in both directions. Thisis what is symbolically shown in FIG. 8, where at the top there is asuccession of pseudorandom sequences C_(E1) and at the bottom asuccessive of pseudorandom sequences C_(E2). Data b₁ impose phase jumpsbetween a sequence C_(E1) and the following sequence C_(E2). Data b₂impose phase jumps between a sequence C_(E2) and the following sequenceC_(E1). The rising or falling oblique arrows symbolize the twoprocesses.

FIGS. 9 and 10 illustrate how it is necessary to modify the diagrams ofthe transmitter and receiver compared with FIGS. 5 and 6 in order toeffect this double transmission. The transmitter of FIG. 9 has twoinputs Ee₁, Ee₂ receiving two binary data flows respectively b₁,k andb₂,k. It comprises two logic circuits 12₁, 12₂ and two delay circuits14₁, 14₂, whose input is connected to the output of one of the logiccircuits and the output is connected to the input of the other logiccircuit. A first multiplier 32₁ processes the symbols d₁,k present atthe output of the logic circuit 12₂ and a second multiplier 32₂ thesymbols d₂,k present at the output of the logic circuit 12₁. There is alocal oscillator 16 supplying the carrier, two modulators 18₁, 18₂ andan adder 31 supplying at its output Se the doubly modulated carrier.

On the receiver side and as is illustrated in FIG. 10, there are twosymmetrical channels crossed at the outset, a first channel with afilter 20₁ matched to the sequence C_(E1) and a delay circuit 22₁ and asecond channel with a second filter 20₂ matched to the sequence C_(E2)and a delay circuit 22₂. Thus, the first channel comprises a multiplier24₁, an integrator 26₁ and a decision logic 28₁. The second channelcomprises a multiplier 24₂, an integrator 26₂ and a decision logic 28₂.The multiplier circuits 24₁, 24₂ receive in the first case the filtered,delayed signal x_(FR1) individual to the sequence C_(E1) and thefiltered signal x_(F2) individual to the sequence C_(E2) and for thesecond the filtered, delayed signal X_(FR2) individual to the sequenceC_(E2) and the filtered signal x_(F1) individual to the sequence C_(E1).These circuits 24₁, 24₂ supply demodulation signals m₁ (t) and m₂ (t),which, as for the simple variant of FIGS. 5 and 6, are at a zero meanvalue outside t=0. The integrators 26₁, 26₂ integrate these signals, asfor FIG. 6, and two logic decision circuits 28₁, 28₂ restore on twooutputs Sr₁, Sr₂, the two data flows b₁,k and b₂,k.

This transmitter-receiver assembly functions as follows. The symbolsd₁,k are the result of the logic combination by exclusive-OR of b₂,k andthe compliment of d₂,k-1. In the same way, the symbols d₂,k are theresult of the logic combination of b₁,k and the compliment of d₁,k-1 andit is therefore possible to write:

    d.sub.1,k =b.sub.2,k ⊕d.sub.2,k-1

    d.sub.2,k =b.sub.1,k ⊕d.sub.1,k-1

The first pseudorandom sequence C_(E1) affects the symbols d₁,k, whereasthe second pseudorandom sequence C_(E2) affects the symbols d₂,k. Thesesequences spread the spectrum of the symbols and new spread spectrumsymbols D₁,k and D₂,k are obtained.

In the receiver, the filter 20₁ of the upper channel, which is matchedto C_(E1), supplies a filtered signal x_(F1), which will depend on thesymbol d₂,k affected by C_(E1), whereas the filter 20₂ of the lowerchannel, which is matched to C_(E2) supplies a signal x_(F2), which willdepend on the symbol d₂,k, which has been affected by C_(E2). However,the delay circuit 22₂ delays the filtered signal x_(F2) in such a waythat the filtered, delayed signal x_(FR2) is dependent on the precedingsymbol, i.e. d₂,k-1, which has been affected by C_(E2). The demodulationperformed by the circuit 24₂ therefore has an effect on a transition ofC_(E2) to C_(E1), which corresponds to the binary data item b₂ of thesecond data flow and in FIG. 8 to one of the rising oblique arrows.

The same reasoning, but symmetrically, shows that the upper channel ofthe demodulator processes the transitions of C_(E1) to C_(E2) andtherefore the data b₁.

The preceding description relates to the transmission of two symbols andthe use of two orthogonal pseudorandom sequences. It is perfectlypossible to extend this principle to n symbols and n sequences, where nis an integer higher than 2. In this case, it is e.g. possible to use atransmission, whose modulation is based on the principle of FIG. 11(compare with FIG. 8) for 3 symbols and 3 sequences.

In this case, the symbol b₁ modulates the phase jumps between a sequenceC_(E1) and the following sequence C_(E2). The symbol b₂ modulates thephase jumps between a sequence C_(E2) and the following sequence C_(E3).The symbol b₃ modulates the phase jumps between a sequence C_(E3) andthe following sequence C_(E1).

All the advantages described hereinbefore are retained. It is merelynecessary for this purpose that the n sequences used are pairwiseorthogonal.

Another embodiment of the principle is illustrated in FIG. 12 and makesit possible to transit, as in the case of FIG. 8, two symbols, but withtwice lower rates and still using two orthogonal pseudorandom sequences.In this case, the two sequences are alternately transmitted and thesymbol b₁ modulates the phase jumps between a sequence C_(E1) and thefollowing sequence C_(E2), whereas the symbol b₂ modulates the phasejumps between a sequence C_(E2) and the following sequence C_(E1). Theadvantage is that due to the fact that at a given instant a singlesequence is transmitted instead of two, i.e. all the transmission poweris used for this sequence, which leads to an improvement in the qualityof the signal received. However, the disadvantage is the reduction ofthe symbol rate.

The construction of the corresponding transmitter is slightly simplifiedcompared with that described relative to FIG. 9, a single modulatorbeing sufficient, there is no adder and the switching of the sequencescan be very easily digitally performed. However, the receiver has astructure strictly identical to that described relative to FIG. 10.

Such a process for the transmission of several bits with the aid ofseveral pseudorandom sequences offers the following advantages comparedwith conventional processes:

a better resistance to scramblers and interference in the wide andnarrow bands, taking account of the passage in the two filters havingorthogonal pulse responses prior to demodulation,

a reduction of the parasitic noise outside the correlation peaks (inmean, effective value).

All the preceding description has related to the transmission of binarysymbols, i.e. a two-state phase modulation. However, the invention isnot limited to this case and is applicable to phase modulations with alarger number of states. The most frequently encountered conventionalconstructions use DQPSK modulation with four phase states permitting thetransmission of two bit symbols. The present invention applies to saidDQPSK modulation. As the phase jumps can assume four possible valuesthey are jumps between two orthogonal sequences instead of being on thesame sequence, as in the conventional case.

The pseudorandom sequences used for spreading the spectrum are notnecessarily binary sequences. The invention can be implemented withsequences having a larger number of states, e.g. quaternary sequences,or even more complex pseudorandom signals.

A particularly interesting case consists of using as a pair oforthogonal pseudorandom sequences, two reciprocal sequences (one beingthe time inverse of the other). Starting from a sequence in C(t), areciprocal sequence is obtained by transforming t into T_(b) -t. The useof reciprocal sequences is e.g. described in WO 92/02997.

This use of two reciprocal sequences make it possible to construct areceiver using a surface acoustic wave component simultaneouslyperforming the matched and delayed filtering operations. A receiverusing such a component is described in FIG. 13.

The component 40 has an input transducer Te and four output transducersT_(s1), T'_(s1), T_(s2), T'_(s2), constituted by aluminium electrodesdeposited on a piezoelectric, e.g. quartz substrate. These fourtransducers are connected to four outputs S₁, S_(r1), S₂,S_(r2).

The four output transducers have an identical pulse response of veryshort duration, equal to or below the duration of one element of thepseudorandom sequences. The global pulse response of the component is,under these conditions, imposed by the geometry of the input transducer.The latter comprises N elementary electrodes in split-finger transducerform, which are interconnected in such a way that the pulse response ofthe filter between the input E and an output S₁ is equal to thetime-reverse combined complex of the first sequence C_(E1). Obviously,if the two output transducers T_(s1) and T_(s2) are arrangedsymmetrically on either side of the input transducer Te, the pulseresponses of the thus formed two filters will be reciprocal of oneanother.

In the same way, by placing two remote transducers T'_(s1) and T'_(s2)on the substrate, in such a way that the distance between T_(s1) andT_(s2) or between T'_(s1) and T'_(s2) is such that the time taken by theacoustic RAYLEIGH wave to cover the interval separating the transducersis equal to T_(b), two delay lines of duration T_(b) will be obtained onthe same component. Thus, this single component 40 can be used in aspectrum spread differential receiver using alternating reciprocalsequences.

It is obviously possible to use two distinct components, each having aninput transducer matched to one of the two orthogonal sequences and twooutput transducers making it possible to obtain filtered signals andfiltered, delayed signals.

I claim:
 1. Process for data transmission by spread-spectrum, comprisingthe steps of:A) on transmission:organizing said data in at least onebinary data sequence (b_(K)), the data having a given period (Tb),coding said data by differential coding to obtain symbols (d_(K)),producing two orthogonal pseudorandom sequences (C_(E1), C_(E2)), eachsequence comprising chips having a duration (Tc), the period (T_(b)) ofsaid data being N times higher than the duration (Tc) of said chips,multiplying said symbols (d_(K)) by said two orthogonal pseudorandomsequences (C_(E1), C_(E2)), so as to obtain two spread symbols (D_(K))₁,(D_(K))₂, generating a carrier, modulating said carrier by said twospread symbols (D_(K))₁, (D_(K))₂, transmitting the modulated carrier,B) on reception:receiving the modulated carrier (x(t)), performing twoparallel filtering operations on the received modulated carrier, saidfiltering operations being respectively matched to said two orthogonalsequences (C_(E1), C_(E2)), said filtering producing two filteredsignals (X_(F1) n X_(F2)), delaying one of said two filtered signals(X_(F)) by said period (T_(b)) of said data and leaving the otherundelayed, estimating the phase difference between said delayed signaland said undelayed signal, restoring said data (b_(K)) from said phasedifference.
 2. Process according to claim 1, wherein said data to betransmitted are organized in a single binary data sequence (b_(K)), saidprocess comprising the steps of:A) on transmission:producing a firstpseudorandom sequence (C_(E1)), multiplying said symbols (d_(K)) by saidfirst pseudorandom sequence (C_(E1)) in order to obtain first spreadspectrum symbols (D_(K))₁, producing a second pseudorandom sequence(C_(E2)) orthogonal to the first (C_(E1)), multiplying said symbols(d_(K)) by said second pseudorandom sequence (C_(E2)) in order to obtainsecond spread spectrum symbols (D_(K))₂, modulating the carrier (P) bysaid first spread spectrum symbols (D_(K))₁ and by said second spreadspectrum symbols (D_(K))₂, B) on reception:receiving the modulatedcarrier (x(t)), performing a first filtering of the signal received(x(t)) by a first matched filter (20₁) having a pulse response which isthe time inverse of the first pseudorandom sequence used on transmission(C_(E1)), this first filtering producing filtered signal (X_(F1)),performing a second filtering of the signal received (x(t)) by a secondmatched filter (20₂) having a pulse response which is the time inverseof the second pseudorandom sequence used on transmission (C_(E2)), thissecond filtering producing a second filtered signal (X_(F2)), delayingone of the filtered signals by one period (T_(b)) to obtain a filtereddelayed signal (X_(FR)), and leaving the other filtered signalundelayed, multiplying said filtered delayed signal (X_(FR)) by saidfiltered undelayed signal (X_(F1)) to obtain a demodulation signal(m(t)), restoring the transmitted data from said demodulation signal. 3.Process according to claim 1, wherein said data to be transmitted areorganized in at least two separate sequences of binary data (b₁,K)(b₂,K), of the same period (T_(b)), said process comprising the stepsof:A) on transmission:forming a first and a second sequences of symbols(d₁,K) (d₂,K) from said two binary data sequences (b₁,K) (b₂,K), onesymbol of a sequence (d₂,K) (d₁,K) being obtained by a logic combinationbetween a data item (b₁,K) (b₂,K), and a symbol of the other sequencedelayed by a period equal to the period (T_(b)) (d₁,K-1, d₂,K-1),producing a first pseudorandom sequence (C_(E1)), multiplying thesymbols of said first sequence (d₁,K) by said first pseudorandomsequence (C_(E1)) to obtain a first sequence of spread spectrum symbols(D₁,K), producing a second pseudorandom sequence (C_(E2)), orthogonal tothe first sequence (C_(E1)), multiplying said symbols of the secondsequence (d₂,K) by said second pseudorandom sequence (C_(E2)) to obtaina second sequence of spread spectrum symbols (D_(K))₂, modulating thecarrier (P) on the one hand by the first sequence of spread spectrumsymbols (D₁,K) and on the other by the second sequence of spreadspectrum symbols (D₂,K), summing the two modulated parts of the carrier,transmitting the carrier thus obtained, B) on reception:receiving themodulated carrier (x(t)), performing a first filtering of said receivedsignal (x(t)) by a first matched filter having a pulse response which isthe time inverse of the first pseudorandom sequence used on transmission(C_(E1)), thus obtaining a first filtered signal (X_(F1)),performing asecond filtering of said received signal (x(t)) by a second matchedfilter having a pulse response which is the time inverse of the secondpseudorandom sequence used on transmission (C_(E2)) thus obtaining asecond filtered signal (X_(F2)), delaying said first filtered signal(X_(F1)) by a period (T_(b)) of the binary data (b₁,K) (b₂,K) and thusobtaining a first filtered delayed signal (X_(FR1)), delaying saidsecond filtered signal (X_(F2)) by a period equal to the period (T_(b))of the binary data (b₁,K) (b₂,K) and thus obtaining a second filtereddelayed signal (X_(FR2)), multiplying said first filtered delayed signal(X_(FR1)) by said second filtered signal (X_(F2)) to obtain a firstdemodulation signal (m₁ (t)), multiplying said second filtered delayedsignal (X_(FR2)) by said first filtered signal (X_(F1)) to obtain asecond demodulation signal (m₂ (t)), integrating said first demodulationsignal (m₁ (t)) for a duration equal to the period (T_(b)) of the binarydata (b₁,K) (b₂,K), integrating said second demodulation signal (m₂ (t))for a duration equal to the period (T_(b)) of the binary data (b₁,K)(b₂,K), evaluating the sign of the signal resulting from the integrationof the first signal and restoring the binary data of the first sequence(b₁,K), evaluating the sign of the signal resulting from the integrationof the second signal and restoring the binary data of the first sequence(b₂,K).
 4. Process according to claim 3, wherein alternate transmissiontakes place of the two orthogonal sequences, a first symbol (b₁)modulating the phase jumps between a first sequence (C_(E1)) and asecond, following sequence (C_(E2)), whereas a second symbol (b₂)modulates the phase jumps between the second sequence (C_(E2)) and thefirst sequence (C_(E1)).
 5. Process according to claim 3, wherein thebinary data are organized in a number n of sequences, n being an integerhigher than 2, n pairwise orthogonal pseudorandom sequences are producedand the data of the two sequences are transmitted with the aid of two ofthe orthogonal pseudorandom sequences.
 6. Process according to claim 1,wherein said symbols (d_(k)) are one bit.
 7. Process according to claim1, wherein said symbols (d_(k)) are two bits.
 8. Process according toclaim 1, wherein said first and second pseudorandom sequences (C_(E1),C_(E2)) are binary sequences.
 9. Process according to claim 1, whereinsaid first and second pseudorandom sequences (C_(E1), C_(E2)) arequaternary sequences.
 10. Process according to claim 1, wherein said twoorthogonal pseudorandom sequences (C_(E1), C_(E2)) are reciprocalsequences.